Reverse energy transfer in zero-current switching power conversion

ABSTRACT

A zero-current switching converter converts power from an input source for delivery to a load. In the converter, a switch opens and closes at times of essentially zero current to enable energy transfer back and forth between said input source and said converter. A controller is connected to control the switch to initiate, during a first portion of each of a succession of converter operating cycles, forward energy transfer from said input source; and in at least some of said converter operating cycles, also to initiate reverse energy transfer to said input source at times not contiguous with said first portion.

This is a divisional of copending application Ser. No. 08/449,698, filedMay 24, 1995.

This invention relates to zero-current switching power conversion.

FIG. 1 shows a circuit schematic of an idealized non-isolated ZCS buckconverter 10. The converter may be operated in either a half-wave or afull-wave mode of operation. In either case, the switch 25 is closed bythe ZCS switch controller 12 at a time of zero flow of current Iin. Inthe half-wave mode the switch 25 is opened by the ZCS switch controller12 on the occurrence of the first zero crossing of current followingswitch closure. Waveforms for the half-wave mode are shown in FIGS. 3Athrough 3D. In the full wave mode the switch 25 is opened by the ZCSswitch controller 12 on the occurrence of the second (or any evennumbered) zero crossing following switch closure. Waveforms for thefull-wave mode are shown in FIG. 4A through 4D. An output filter 16 inthe converter 10 filters the pulsating voltage Vc(t) to deliver anessentially DC output voltage, Vout, across the load. In the converter10, a frequency controller 15 compares the converter output voltage,Vout, to a reference value (indicative of some desired value of Vout,but not shown) and sets the converter operating frequency, fop, to avalue consistent with maintaining Vout at the desired value. Thefrequency fop, delivered to the ZCS switch controller 12, controls therate of occurrence of operating cycles (e.g., fop=1/(t5-t0) in FIGS. 2,3 and 4). In general, a ZCS converter will exhibit higher conversionefficiency than a conventional pulse-width modulated ("PWM") buckconverter (20, FIG. 2) because opening and closing the switch 25 atessentially zero current results in a substantial reduction in switchinglosses. The reduction in switching losses allows for an increase inconverter operating frequency and a corresponding decrease in the sizeof a ZCS converter relative to its PWM counterpart.

In half-wave mode, a finite and bounded amount of energy is withdrawn bythe converter from the input source 14 during each converter operatingcycle (the "half-wave" energy) and all of this energy is delivered tothe load during the cycle. As a result, maintaining a constant loadvoltage requires that converter operating frequency be variedessentially in proportion to the converter load. If the converter isoperated in full-wave mode, the first half of the operating cycle isidentical to the half-wave operating mode and an amount of energy equalto the half-wave energy is withdrawn from the input source 14. However,during the second half of the operating cycle (between the first andsecond zero crossings) a portion of this energy is returned to the inputsource 14 with the remainder being delivered to the load. In thefull-wave mode, the ratio of the amount of the energy which is returnedto the source 14 to the amount of the energy which is provided to theload varies with converter loading; as a result, converter operatingfrequency tends to be relatively constant and independent of load. Thehalf-wave ZCS converter is a "quantized" converter in the sense that apredictable amount of energy is delivered to the load during eachconverter operating cycle. This is in contrast to the full-wave ZCSconverter in which a variable fraction of the total half-wave energyextracted from the input source is actually delivered to the load duringeach operating cycle.

Processing of energy in the half-wave mode is potentially more efficientthan in full-wave mode since all of the energy which is delivered to theconverter during each operating cycle is transferred forward to the loadwithout recirculation. In practical converter embodiments, losses incircuit elements occur as result of both the forward and the reverseflow of energy. In the half-wave converter there is no reverse energyflow and the reduction in operating frequency with decreasing loadresults in conversion efficiency remaining relatively high over a widerange of loads. In the full-wave converter, the combination of reverseenergy flow and a relatively constant, high, operating frequencymitigate toward a reduction in conversion efficiency as load is reduced.Another benefit of the quantized nature of half-wave conversion is thathalf-wave ZCS converters operating in synchronism will accurately sharein the power delivered to a load; practical embodiments of full-waveconverters will not. On the other hand, the relatively constant, high,operating frequency of the full-wave mode mitigates toward both asmaller converter (e.g., the breakpoint frequency of the converteroutput filter may be raised, thereby reducing the size of the filtercomponents) and improved converter performance (e.g., the higher outputfilter breakpoint frequency will allow wider overall converter bandwidthand a concomitant improvement in transient response).

Examination of the waveforms of FIGS. 3 and 4 show that the switch in ahalf-wave ZCS converter must be able to carry a unipolar current and becapable of blocking a bipolar voltage, whereas the switch in thefull-wave converter must carry a bipolar current and block a unipolarvoltage. As a practical matter, semiconductor switching elements whichcan block a bipolar voltage when off and which exhibit switching speedsconsistent with efficient converter operation at relatively highoperating frequencies (e.g. 500 KHz or 1 MHZ) are not currentlyavailable. For example, FIG. 5A shows a ZCS buck converter 30 using aMOSFET transistor 26 as the switching element. Such a device is notcapable of supporting a bipolar voltage due to the "intrinsic" diode 31inherent to the MOSFET. As a result, while the converter embodiment ofFIG. 5A may be used in the full-wave operating mode (the MOSFET switchis turned on to conduct the positive current flow of Iin; the diode willconduct the negative flow; the switch need only block a unipolarvoltage) it cannot be used in the half-cycle mode. A ZCS converter 30,adapted to operate in half cycle mode and shown in FIG. 5B, includes acomposite switch 125, comprising a diode 38 in series with a MOSFETswitch 26, capable of blocking negative values of Vsw. The lossesattendant to the flow of current Iin in this diode results in areduction in conversion efficiency. Similar considerations apply if abipolar switch is used in a ZCS converter: in a half-wave converter aseries diode (38, FIG. 6A) will generally be needed to form a compositeswitch 25a capable of blocking negative values of Vsw; in a full-waveconverter an anti-parallel diode (34, FIG. 6B) is required to conductnegative flow of Iin.

While the preceding discussion has been directed to a particular ZCSconversion topology (the non-isolated buck converter) it is moregenerally applicable to a wide variety of ZCS conversion topologies. Forexample, both isolated and non-isolated ZCS converters of the buck,boost, buck-boost, forward, flyback, Cuk, transformer coupled Cuk andSEPIC topologies are described in Liu, et al, "Resonant Switches--AUnified Approach to Improve Performances of Switching Converters," IEEEInternational Telecommunications Conference, 1984 Proceedings, pp.344-351; Liu, et al, "Resonant Switches--Topologies andCharacteristics," PESC '85 Record, 16th Annual IEEE Power ElectronicsSpecialists Conference, pp. 106-16; in Vinciarelli, et al, "BoostSwitching Power Conversion," U.S. Pat. No. 5,321,348, Jun. 14, 1994; andin Lee, et al, "Zero-Current Switching Quasi-Resonant ConvertersOperating in a Full-Wave Mode," U.S. Pat. No. 4,720,667, Jan. 19, 1988(all incorporated herein by reference). The references illustrate theuse of diodes in series or in parallel with MOSFET and bipolar switchesto adapt the switches to half-wave or full-wave operation and alsodiscuss the concept of the "resonant switch" as applied to ZCSconversion. In general, the resonant switch concept provides forreplacing a switching element in a PWM converter topology with aresonant switch comprising a switching element, a capacitor, an inductorand one or more diodes (depending on whether the application is ahalf-wave or full-wave) to create a corresponding ZCS topology.

Means for limiting the minimum operating frequency of a half-wave ZCSconverter are described in Vinciarelli, et al, "Zero Current SwitchingForward Power Conversion With Controllable Energy Transfer," U.S. Pat.No. 5,235,502, Aug. 10, 1993; in Vinciarelli, "Zero Current SwitchingForward Power Conversion Operating in Damped Reverse Boost Mode," U.S.Pat. No. 5,291,385, Mar. 1, 1994; and in Vinciarelli, "Power Conversionin Anticipatory Reverse Boost Mode," U.S. patent application Ser. No.08/187,296, filed Jan. 27,1994 (all incorporated herein by reference).Each of the referenced techniques exploits the fact that there is avalue of converter loading below which there will be a reversal of theflow of current in the output inductor 50 during a portion of theoperating cycle. By adding a second switch 44 in parallel with theresonant capacitor 42 (FIG. 7), and timing the opening and the closingof the second switch 44 to control the charging of the capacitor 42 viathe reverse flow of Io, the amount of energy transferred forward duringeach operating cycle can be reduced, thereby requiring a higherconverter operating frequency than that which would otherwise berequired to transfer the particular amount of power to the load.

Saturable inductors are well known. They typically consist of a windingon a magnetic core having a known value of saturation flux density(where the "saturation flux density" is the value of flux density belowwhich the permeability of the core is relatively high and above whichthe value of the permeability of the core is relatively low), exhibit arelatively high value of inductance at relatively low values of current(e.g. values which do not cause the flux density in the core to exceedthe saturation flux density) and a relatively much lower value ofinductance at relatively higher currents (e.g. values which wouldotherwise cause the flux density in the core to exceed the saturationflux density). Viewed another way; if a voltage is applied across asaturable inductor the flux in the core which links the winding, at anyinstant in time, will be proportional to the time integral of thevoltage across the winding (i.e. the "volt-seconds" applied to thedevice). If the applied "volt-seconds" exceeds a certain value(dependent upon the number of turns in the winding and the dimensionsand saturation flux density of the core) the magnetic flux linking thecore will exceed the saturation flux density and the inductance of thedevice will drop.

Using a saturable inductor to modify the shape of the current andvoltage waveforms in both half-wave and full-wave ZCS converters, ofboth the buck and boost types, is described in Erikson, "NonlinearResonant Switch and Converter," U.S. Pat. No. 4,829,232, May 9, 1989. InErikson's converter the saturable inductor is used in place of a fixedresonant inductor (e.g., inductor 27, FIG. 1) and arranged to saturateand assume a low inductance value at low values of current and tounsaturate and assume a high inductance value at high currents. Thiscauses the characteristic time constant for the converter to increasewith increasing switch current, resulting in a current waveform whichexhibits relatively fast rise and fall times separated by a longer,relatively flat, central region. This reduce the rms value of thecurrent which flows in the switch (and in the series diode in thehalf-wave converter), thereby improving converter efficiency.

Using saturable inductors in combination with a switching element havingunipolar voltage blocking capability to provide a composite switchcapable of blocking a time-varying voltage which exhibits a briefreversal in polarity is described in Vinciarelli, et al, "BoostSwitching Power Conversion Using Saturable Inductors," U.S. patentapplication Ser. No. 07/887,486 (incorporated by reference). Thatapplication describes a ZCS boost converter topology which, asillustrated in FIG. 8, exhibits a bipolar capacitor voltage, Vc(t), whenoperated in the short-cycle operating mode. Ordinarily, the switch 39would have to be able to withstand a bipolar voltage for a period oftime following the opening of the switch (e.g., between times ta ant tbin the waveform). One way to achieve this would be to place a diode inseries with the switch 39; another way, described in that application,is to connect a saturable inductor 37, having a volt-second capabilitysufficiently large to support the voltage Vc(t) throughout the time thatthe voltage is negative (e.g., between times ta and tb) subsequent tothe switch being opened, in series with the switch 39. A general classof ZCS boost converters using coupled inductors, and semiconductorswitch topologies for use in both half-wave and full-wave versions ofthe converters, are described in Vinciarelli, et al, "Boost SwitchingPower Conversion," U.S. Pat. No. 5,321,348, Jun. 14, 1994 (incorporatedby reference).

ZCS boost converters which deliver a DC output but which are operateddirectly from an AC (e.g., bipolar) input source are described inVinciarelli, et al, "AC to DC Boost Power Converters," U.S. patentapplication Ser. No. 08/274,991, filed Jul. 13, 1994 (incorporated byreference).

SUMMARY OF THE INVENTION

In general, in one aspect, the invention features a zero-currentswitching converter for converting power from an input source fordelivery to a load. In the converter, a switch opens and closes at timesof essentially zero current to enable energy transfer back and forthbetween said input source and said converter. A controller is connectedto control the switch to initiate, during a first portion of each of asuccession of converter operating cycles, forward energy transfer fromsaid input source; and in at least some of said converter operatingcycles, also to initiate reverse energy transfer to said input source attimes not contiguous with said first portion.

Implementations of the invention may include one or more of thefollowing features. The forward energy transfer may be initiated byclosing the switch. The reverse energy transfer may also be initiated byclosing said switch. The initiation of reverse energy transfer may occura predetermined time after the beginning of the first portion. Theminimum frequency of the converter operating cycles may be controlled bycontrolling a time delay between the beginning of the first portion andthe initiation of the reverse energy transfer. There may be circuitelements (e.g., an inductor and a capacitor) which set a characteristictime constant for the rise and fall of a flow of current after theswitch is closed. There may be a saturable inductor and the time betweenthe beginning of the first portion and the initiation of reverse energytransfer may be determined by the volt-second characteristic of thesaturable inductor. Energy may be transferred forward from the inputsource to the capacitor during the first portion and may be transferredfrom the capacitor back to the input source during the reverse energytransfer. The switch may include two switching elements each connectedto transfer energy between the input source and the converter. A firstswitching element, connected in series with a first (e.g., saturable)inductor, may set a characteristic time constant associated transfer ofenergy from the input source. A second switching element, connected inseries with a second (e.g., saturable) inductor, may set acharacteristic time constant associated with the transfer of energytoward the input source. A saturable inductor may be connected to carrythe current (currents) which flows in the switching element (elements).

The zero-current switching converter may be a buck converter, a boostconverter, and AC-DC boost converter, a buck-boost converter, a Cukconverter a SEPIC converter, a non-isolated converter, or an isolatedconverter, among others.

There may be circuitry for reducing the volt-seconds required to bringthe inductor from an unsaturated to a saturated state. The input sourcemay include a unipolar voltage. The switch may be connected in serieswith the input source and the first inductor. There may be an outputfilter connected between the capacitor and the load for deliveringenergy from the capacitor to the load at an average voltage essentiallyequal to the average voltage across the capacitor. A diode may beconnected in parallel with the capacitor and poled to block a voltage ofthe polarity of the average voltage. An input inductor may be connectedin series with the input source and the series circuit, and a diode maybe connected between the capacitor and the load, the diode poled toconduct current in a direction to deliver energy from the input sourceto the load. The switching element may be a unidirectional switch poledto block the voltage delivered by the input source when the switchingelement is open. The saturable inductor may be arranged to exhibit arelatively high unsaturated impedance during portions of the operatingcycle during which reverse voltage would otherwise have to be blocked bythe switching element. The input source may be a bipolar voltage source,the capacitor may be connected in parallel with the series circuitcomprising the switch and the first inductor, an input inductor may beconnected in series with the input source and the series circuit, and arectifier may be connected between the dual-mode switch and the load,the rectifier accepting a bipolar input current and delivering aunipolar output current to the load. The inductance may be the leakageinductance of a leakage-inductance transformer comprising a primarywinding and a secondary winding. The first switching element may beconnected in series with the primary winding. A second switching elementmay be connected in series with both the secondary winding and thecapacitor, the first switching element and the primary winding beingconnected in series with the input source. An output filter may beconnected between the capacitor and the load for delivering energy fromthe capacitor to the load at an average voltage essentially equal to theaverage voltage across the capacitor. A diode may be connected inparallel with the capacitor and poled to block a voltage of the polarityof the average voltage.

The reverse energy transfer may be initiated if the voltage across thecapacitor is greater than the voltage delivered by the input source. Thereverse energy transfer may be initiated if the voltage across thecapacitor is at a value such that the absolute value of the incrementalchange in the capacitor voltage which must occur to enable current toflow into the load is greater than the voltage across the load.

In general, in another aspect, the invention features a method ofcontrolling the minimum operating frequency of a zero-current switchingconverter which transfers power from an input source for delivery to atime-varying load by opening and closing a switch at times of zerocurrent flow. In a first portion of each of a series of converteroperating cycles, the switch is controlled to initiate forward energytransfer from the input source. In response to changes of the load,during each of some of the converter operating cycles and at times whichare not contiguous with the first portion, controlling the switch toinitiate reverse energy transfer to the input source.

Implementations of the invention may include one or more of thefollowing features. A fixed time delay may be caused between thebeginning of the first portion and the initiation of the reverse energytransfer. The time delay between the beginning of the first portion andthe initiation of the reverse energy transfer may be selectivelyadjusted. The volt-second characteristic of a saturable inductor may beadapted to effect a controlled time delay between the beginning of thefirst portion and the initiation of the reverse energy transfer. Thereverse energy transfer may be initiated if doing so will result inenergy being transferred from the converter back to the input source.The reverse energy transfer may be initiated on the basis of a voltagemeasured across a capacitor.

Other features and advantages of the invention will become apparent fromthe following description and from the claims.

DESCRIPTION

FIG. 1 shows a prior art non-isolated zero-current switching buckconverter.

FIG. 2 shows a prior art non-isolated pulse-width modulated buckconverter.

FIGS. 3A-3D show waveforms for the ZCS converter of FIG. 1 when operatedin the half-wave mode of operation.

FIGS. 4A-4D show waveforms for the ZCS converter of FIG. 1 when operatedin the full-wave mode of operation.

FIGS. 5A and 5B show prior art embodiments of ZCS buck converters, usingMOSFET semiconductor switches, adapted for operation in the full-waveand half-wave modes of operation, respectively.

FIGS. 6A and 6B show composite switches comprising bipolar transistors.

FIG. 7 shows a ZCS converter which incorporates a second switch whichcan be used to limit the minimum operating frequency of the converter.

FIG. 8 shows a prior art ZCS boost converter which includes a saturableinductor in series with the switch.

FIGS. 9A through 9D show capacitor voltage waveforms for the converterof FIG. 1 at different values of load.

FIG. 10 shows a schematic of a ZCS buck converter according to theinvention.

FIGS. 11A and FIGS. 11B and 11C show, respectively, a functionalschematic of a multi-mode switch and two dual-mode switch controllers.

FIGS. 12A through 12D show waveforms for the converter of FIG. 10.

FIGS. 13A through 13D show capacitor voltage waveforms for the converterof FIG. 10 at different values of load.

FIG. 14 shows a multi-mode switch using a single inductor.

FIG. 15A and FIGS. 15B through 15E show a circuit symbol used toindicate a unidirectional switch and several embodiments ofunidirectional switches, respectively.

FIGS. 16A through 16D show embodiments of multi-mode switches.

FIG. 17 shows a portion of a ZCS converter which includes a multi-modeswitch comprising a saturable inductor.

FIG. 18 shows a magnetization characteristic of a saturable inductor.

FIGS. 19A and 19B show waveforms for the converter of FIG. 17.

FIGS. 20A through 20D show waveforms for the converter of FIG. 17 atdifferent values of load.

FIGS. 21A and 21B show additional operating waveforms for the converterof FIG. 17.

FIGS. 22A and 22B show capacitor voltage waveforms for the converter ofFIG. 17 at different values of load.

FIGS. 23A and 23B show means for resetting the saturable inductor inconverters.

FIG. 24 shows another magnetization characteristic of a saturableinductor.

FIG. 25 shows another embodiment of a multi-mode switch comprising asaturable inductor.

FIGS. 26A and 26B show a circuit schematic and waveforms, respectively,for a prior art ZCS buck converter.

FIGS. 27A and FIGS. 27B through 27D show, respectively, a circuitschematic and operating waveforms for a ZCS buck converter.

FIG. 28 shows a prior art AC-DC ZCS boost converter.

FIG. 29 shows an AC-DC ZCS boost converter.

FIGS. 30A and 30B show bipolar multi-mode switches for use inconverters.

FIGS. 31A and 31B show schematics of a prior art ZCS forward converterand its dual-mode ZCS counterpart, respectively.

FIG. 32 shows a portion of an embodiment of the converter of FIG. 31Busing MOSFET semiconductor switches.

FIGS. 33A and 33B show M-Type and L-Type dual-mode resonant switches,respectively.

Waveforms of the capacitor voltage, Vc(t), for a prior art ZCS buckconverter of the kind shown FIG. 1, operating in half-wave mode at aconstant output voltage Vout, at three different values of load power,are shown in FIGS. 9A through 9C. For simplicity we assume that thevalue of Lo is very much larger than Lr1, so that the value of Io isessentially constant throughout an operating cycle. At maximum load,Pout=Pmax (for which Io=Ip), the converter operating period is t1, asillustrated in FIG. 9A; for a second, lower, value of load,Pout=P1<Pmax, the period is t2>t1, as shown in FIG. 9B; for third andfourth value of loads, P2 and P3, where P3<P2<P1, the periods are t3 andt4, respectively, as shown in FIGS. 9C and 9D. In all cases, assuming alossless output filter 16, the period, and hence the operatingfrequency, will assume a value such that the average value of Vc(t) overthe operating cycle is equal to Vout. While circuit losses in physicallyrealizable ZCS converters will set a finite lower limit on minimumconverter operating frequency, the theoretical operating frequency of anunloaded converter will be zero. There are benefits to putting apredictable lower limit on operating frequency: the output filterbreakpoint frequency may be raised which mitigates toward smaller, lesslossy, filter components and wider converter bandwidth; conducted andradiated EMI/RFI filters and shields become smaller; and the potentialfor frequency-specific interference within load circuitry can beeliminated.

A circuit model which demonstrates the operating principle of a ZCSconverter topology which provides for setting a lower limit on converteroperating frequency while retaining the benefits of half-wave operation(e.g., efficiency, power sharing) at elevated loads, is illustrated inFIG. 10. In the Figure a ZCS buck converter 100 includes the samecircuit elements as the prior art ZCS buck converter of FIG. 1, exceptthat the switch 25 and the resonant inductor 27 of FIG. 1 are replacedwith a "multi-mode switch" 105 in the converter of FIG. 10. Themulti-mode switch consists of a first switching element 75a in serieswith a first resonant inductor 27a, of value Lr1, and a second switchingelement 75b in series with a second resonant inductor 27b, of value Lr2.The multi-mode switch 105 also includes a dual-mode switch controller112, a simplified functional schematic of which is shown in FIG. 11A.With reference to FIGS. 10, 11A and 12, the dual-mode switch controller105 receives a series of trigger signals 77 at a converter operatingfrequency fop=1/top (FIG. 12). Upon receipt of each trigger signal, thefirst ZCS switch controller 12a opens and closes the first switch 75a atzero current to initiate a forward flow of current, Iin (FIG. 12C), andforward transfer of energy from the input source 14 toward the capacitor42 via the first inductor 27a. The sinusoidal variations in the currentsand voltages in the converter during this forward energy transfer phaseexhibit a characteristic time constant equal to T1=pi*sqrt(Lr1*Cr). Thisinitial part of the operating cycle is seen to be essentially identicalto the initial portion of an operating cycle in the prior art ZCS buckconverter operating in half-wave mode (e.g., as shown in FIG. 3).However, if a new trigger signal is not received within a predetermineddelay time, Td, after the initiation of forward energy transfer (whereTd ends after the forward energy transfer phase has ended, e.g. at atime Td>t1 in FIG. 12), the second ZCS switch controller 12b will betriggered by the output of delay element 17 causing switch 75b to openand close at zero current, and resulting in a reverse flow of current,Iin, and a reverse flow of energy from the capacitor 42 back to theinput source 14 via inductor 27b (provided that the capacitor voltage,Vc(Td) is greater than the input voltage, Vin, at time Td; if thevoltage Vc(Td) is below Vin, the second switch will not be activated).The sinusoidal variations in the currents and voltages in the converterduring this reverse energy transfer phase exhibit a characteristic timeconstant equal to T2=pi*sqrt(Lr2*Cr).

If, as illustrated in FIG. 12D, the voltage Vc(Td) is above Vin by anamount V1, then the voltage Vc will ring down in an amount 2* V1 betweentimes Td and t2 during the reverse energy transfer phase. As load isreduced, the value of V1 will increase toward Vin and Vc(t2) willdecrease toward zero. If the value of Io is assumed to be essentiallyconstant throughout the operating cycle, then the rate-of-change ofVc(t) between times t1 and Td and between times t2 and t3 will be thesame (and equal to dVc/dt=Io/Cr) and hence the two periods, Ts=Td-t1 andTs=t3-t2, will also be the same. It is also to be noted that the totalarea under the voltage waveform of FIG. 12D, during the two dischargeperiods labeled Ts, is equal to Ts*Vpk. If we make the approximatingassumption that this represents the bulk of the area under the waveform,then the area under the waveform will be essentially independent ofload, as will the converter operating frequency to maintain a constantvalue of Vout. A comparison of FIGS. 13A through 13D, which showwaveforms for Vc(t) in the converter of FIG. 10, with FIGS. 9A through9D, which show waveforms for Vc(t) in the converter of FIG. 1, can beused to illustrate the principle. As discussed above, the waveforms ofFIG. 9 illustrate that a significant reduction in operating frequency isrequired in a prior art ZCS converter operating in the half-waveoperating mode to maintain a constant output voltage as load drops. If,however, as illustrated in FIGS. 13A through 13D, the half-wave cycle isaborted at time Td and energy stored in the capacitor 42 is returnedback to the input source, then the variation in operating frequency canbe substantially reduced. In FIGS. 9 and 13, the characteristic timeconstant for forward energy transfer, T1 =pi,sqrt(Lr1*Cr), is assumed tobe the same and the output power, Pout, for each corresponding waveformis also the same. The waveform in FIG. 9A is the same as that in FIG.13A and the waveforms in FIGS. 9B and 13B are also the same, because, inboth cases, Vc is below Vin at time Td; reverse energy transfer will notoccur for Vc<Vin and is therefore not initiated. However, in FIGS. 13Cand 13D the voltage Vc is greater than Vin at time t=Td; as a result ofreverse energy flow the variation in converter operating frequency isreduced substantially compared to the prior art half-cycle converter.

As illustrated in FIG. 12 and FIGS. 13A-13D, the switch 105 will operatein different modes depending upon converter operating conditions. Atconditions of elevated load (e.g., FIGS. 13A and 13B) the switch willoperate only in the half-wave mode and Iin will flow only in the forwarddirection toward the capacitor 42. As loading is reduced, however, theswitch 105 will operate in a mode such that both forward and reverseflow of Iin are caused to occur. Unlike a conventional full-waveconverter, however, in which reversal of the current Iin beginsimmediately following the end of forward current flow (e.g., at timet=tr, FIG. 4A), reverse current flow in the present converter does notbegin until a finite time has elapsed after forward current flow hasended. By providing for half-wave operation from maximum converter loaddown to a preselected lower value of load the efficiency and powersharing benefits of the half-wave converter can be provided atrelatively high values of load, where they are of the most benefit. Atlower values of load, initiation of reverse energy transfer puts a lowerbound on converter operating frequency and provides for the previouslydescribed benefits.

A more detailed schematic of a dual-mode controller 112, of the kindshown in FIG. 11A, is shown in FIG. 11B. In the Figure, trigger signals77 arriving at the operating frequency fop are delivered both to firstZCS switch controller 12A and to retriggerable monostable 117. The firstZCS controller initiates forward half-wave flow of current Iin.Following each trigger signal, the retriggerable monostable 117generates a logically high output for a time period equal to Td (and, iffop exceeds 1/Td, the output of the monostable will stay in a logicallyhigh state). Inverter 72 and AND gate 74 combine the output of themonostable 117 with the output of comparator 70 (a signal indicative ofwhether the capacitor voltage Vc is greater than the converter inputvoltage Vin). If the time period Td has elapsed and Vc>Vin, then thesecond monostable 76 will be triggered to produce a brief pulse foractivating the second ZCS controller 126 and initiating a reverse energytransfer phase. If Vc<Vin at time Td, or if fop>1/Td, the second ZCScontroller will not be activated.

The converter 100 and multi-mode switch 105 of FIGS. 10 and 11 providefor two distinct characteristic time constants for the forward andreverse energy transfer phases. In many applications this will not be arequirement and the multi-mode switch 105 may be simplified to the formshown in FIG. 14. In the Figure, the multi-mode switch 105 uses a singleinductor 27 to provide a single characteristic time constant for boththe forward and reverse energy transfer phases, and a dual-modecontroller 112 for opening and closing the single ideal switch 75 (whichis able to block a bipolar voltage when open and carry a bipolar currentwhen closed) at the same times that the dual-mode controller 112 openedthe two ideal switches 75a, 75b of FIG. 11.

The converter and multi-mode switch of FIGS. 10 and 11 have beendescribed and illustrated using ideal switches. In practice, non-idealswitches will be used and, as discussed earlier, application of theseswitches must take into account their limitations. Since a number ofdifferent types of semiconductor devices, such as bipolar and MOSFETswitches, can be applied, it will be convenient to use the symbol 130 inFIG. 15A to indicate a unidirectional switch, e.g., a switch which iscapable of blocking a voltage, when open, in a direction indicated bythe positive and negative polarity marks included in the switch symbol,but which is either incapable, or has very limited capability ofblocking a voltage of the opposite polarity. Thus, for example, thegeneralized unidirectional switch 130 of FIG. 15A is capable of blockinga voltage Vsw having the polarity indicated in the Figure and istherefore symbolic of the N-channel MOSFET 61 poled as shown in FIG. 15Band the P-channel MOSFET 62 poled as shown in FIG. 15C. It is alsosymbolic of the NPN transistor 63 poled as shown in FIG. 15D and the PNPtransistor 64 poled as shown in FIG. 15C. It should be noted, however,that while the intrinsic diodes 31 inherent to MOSFETs 61, 62 make thesedevices virtually incapable of supporting any reverse voltage, thebipolar transistors 63, 64 can typically support a low value of reversevoltage (e.g., negative values of Vsw up to about 6.5 volts, which istypical of the reverse avalanche breakdown voltage rating of thebase-emitter junction of a bipolar transistor). In general, however, thebipolar devices may be considered to be unidirectional switches.

FIGS. 16A through 16D show embodiments of multi-mode switches 105 whichcomprise unidirectional switches 105 (using the symbol defined in FIG.15A). The switches of FIGS. 16A and 16C include two inductors to providetwo different characteristic time constants for the forward and reverseenergy transfer phases. The switches are also assumed to include adual-mode controller, which is not shown in the Figures. Like the switch125 in the half-wave prior art converter 30 of FIG. 5B, the compositeswitch configurations of FIG. 16 suffer the efficiency penalty,previously described, inherent to the presence of a series diode in theforward current path (e.g., diode 38a, FIG. 16A and diode 134b, FIG.16D). The switch configurations of FIGS. 16A and 16C use two inductors27a, 27b to provide two different characteristic time constants for theforward and reverse flow of current; the switch configurations of FIGS.16B and 16D include a single inductor 27 to provide a singlecharacteristic time constant.

A functional schematic of a multi-mode switch which overcomes theefficiency penalty associated with use of a series blocking diode isillustrated in FIG. 17. The multi-mode switch 200 consists of aunidirectional switch 130 in series with both a saturable inductor 250and a linear inductor 227 of value Lr1 (a "linear inductor" has aninductance value which is fixed and independent of the value of Iin overthe expected range of variation of Iin); the saturable inductor has asaturated inductance, Lsat, which is much smaller than Lr1 and anunsaturated inductance, Lunsat, which is much greater than Lr1. For abipolar symmetrical excitation, the series combination of the saturableinductor 250 and the linear inductor 227 might have a compositemagnetization curve of the kind shown in FIG. 18. In the Figure, if thecore is initially unenergized (e.g., in FIG. 18, at the point marked "a") a positive flow of current Iin will bias the core along the pathindicated by the arrow 212. This is a region in which the saturableinductor is saturated in the forward direction and the compositeinductance is essentially equal to Lr1. As current is reduced (e.g.,along paths 214 and 216), however, the saturable inductor will come outof saturation and the composite inductance will increase to a valueessentially equal to Lunsat. As indicated in the Figure, the slope ofthe magnetization curve is indicative of the composite inductance. The"volt-second rating" of the saturable inductor is the total volt-secondswhich the inductor can support while in its unsaturated state (e.g., thetime integral of the inductor voltage, Vsat, required to cause the fluxto change from forward saturation (e.g., point "c", FIG. 18) to reversesaturation (e.g., point "d", FIG. 18) along a path like that of path 219in the magnetization curve of FIG. 18). The current levels at which thesaturable inductor makes transitions between its saturated andunsaturated states are assumed to be small relative to the peak value ofIin which flows during a forward energy transfer cycle.

The operating principle of the multi-mode switch 200 of FIG. 17 isexplained with reference to FIGS. 17 and 18 and the waveforms of FIGS.19. At time t=t0, the unidirectional switch 130 is turned on (at zerocurrent) with the saturable inductor 250 initially in a saturated state(e.g., at a point like point "a" 210 in FIG. 18). As indicated in FIG.19A, there will be an initial half-wave flow of current, Iin, betweentimes to and t3. Between times te and tx, the voltage Vc(t) (FIG. 19B)will be above Vin, inducing a reversal of the flow of Iin back towardthe source (the unidirectional switch 130 being incapable of blockingthe reverse flow). This will bias the saturable inductor 250 into anunsaturated state (e.g., along a path like path 216, FIG. 18) causingthe composite inductance to rise sharply (e.g., to a value Lunsat).Because of the relatively high impedance presented by Lunsat, both thereverse current which flows after time t3, and the amount of energyreturned back to the input source 14, will be very small. Between timest3 and tc, the saturable inductor 250 performs the function performed bythe series diode 38 in a prior art half-wave ZCS converter, supportingVsw and preventing substantial reverse current from flowing during thetime that Vsw is negative. At time t=tc, however, the volt-secondssupported by the saturable inductor (e.g., as represented by the shadedarea 221 under the waveform for Vc(t) in FIG. 20B) exceeds theinductor's volt-second rating; the saturable inductor becomes reversesaturated, the composite inductance drops back to Lr1, and a reverseflow of Iin between time tc and tr transfers a portion of the energystored in the capacitor 42 back toward the input source 18. Here themagnetization curve would be traversed along a path like that of 219 inFIG. 18, returning back to zero current (e.g., to point "e", FIG. 18)along a path like that of path 220. As discussed below, the saturableinductor must be reset to forward saturation to initiate another forwardenergy transfer cycle.

As load is increased, the rate of decline of Vc(t) will increase and,above some value of load power, Px, the net volt-seconds applied to theinductor while Vc is above Vin will no longer be sufficient to bring theinductor into reverse saturation. For example, as illustrated in FIGS.20A and 20B, if the converter is operating at its maximum allowable load(e.g., at a load Pmax, for which Io=Ip), the current Iin will return tozero at the same instant that Vc has declined to a value Vin. For thisoperating condition the saturable inductor is not called upon to supportany reverse voltage between times t3 and tx; the saturable inductor willremain in forward saturation (e.g., at point "a", FIG. 18) and will nothave to be reset prior to initiation of another forward energy transfercycle. On the other hand, as illustrated in FIGS. 20C and 20D, as loaddeclines from Pmax toward Px the saturable inductor will be called uponto support reverse voltage for an increasing period of time, tx--t3, asindicated by the cross-hatched area 223. In general, as load isdecreased from Pmax toward Px, the volt-seconds required to reset thecore back to forward saturation will increase.

One way to reset the saturable inductor to forward saturation is tosimply turn on the unidirectional switch 130. Assume, for example, thatthe saturable inductor is in reverse saturation and that a new operatingcycle is started. Closure of the switch 130 (FIG. 17) will result inapplication of Vin across the composite inductor; the saturable inductor250 will become unsaturated as it begins to traverse a path like that ofpath 225 in FIG. 18 and, as shown in FIG. 21A, the application of Vinacross the unsaturated inductor will result in a small positive flow ofcurrent Iin between times to and tf. The volt-seconds associated withthe application of Vin across the saturable inductor will result in theinductor going into forward saturation at time t=tf and initiation of ahalf-wave forward flow of current between times tf and t3. The rest ofthe operating cycle will be as described earlier. Using Vin to reset thesaturable inductor to forward saturation at the beginning of eachoperating cycle effectively results in a delay in the initiation inforward energy transfer and a concomitant increase the length of eachcycle. To estimate the relative size of this delay, we again note thatthe volt-second rating of the saturable inductor is indicated by thearea under the Vc(t) waveform between times t=t3 and t=tc. This area isillustrated for two different values of converter load in FIGS. 22A and22B. In FIG. 22A the converter is at zero load: Vc rings up to 2*Vin andremains there until the saturable inductor saturates. If the saturableinductor has a volt-second rating equal to VS, then the time period Ts1will be Ts1=VS/Vin. In FIG. 22B, all other converter parameters remainthe same except that the load is reduced to Px, the maximum value ofload at which reverse saturation will still occur (e.g., the saturableinductor will saturate at the exact time that the voltage Vc hasdischarged to a value Vin). The time period Ts2 in FIG. 22B will betwice Ts1, since the volt-seconds associated with each of the areas221a, 223b, and hence the two areas themselves, must be the same. Thus,the value of Ts will vary over an approximately 2 to 1 range asconverter load is increased from no-load to Px. However, resetting ofthe saturable inductor back to positive saturation always occurs at aconstant value of voltage, Vin, so the delay time, Treset (FIG. 21) willbe approximately equal to Treset=VS/Vin. Therefore, the delay timeTreset will be comparable in magnitude to Ts1 (and approximatelyone-half of the magnitude of Ts2). As load is increased above Px, andthe volt-seconds required to reset the core decreases, the delay timeTreset will decline toward zero.

Other ways to reset the saturable inductor, which will also reduce oreliminate the delay time of FIG. 22, are illustrated in FIG. 23A and23B. In FIG. 23A, for example, the multi-mode switch of FIG. 17 ismodified by the addition of a diode 238. Use of this diode is generallybeneficial as it will help control voltages and current flows when theunidirectional switch 130 is turned off. For example, if the switch 130is turned off when a small positive current Iin is flowing, the diode238 will provide a path for the flow of the current and clamp thevoltage Vs at ground, thereby providing path for the flow of Iin towardthe capacitor 42 and protecting the switch 130 from exposure touncontrolled negative transient voltages. However, if the diode is alsosized to exhibit a lower voltage drop than the catch diode 46, then aportion of the current Iout can flow in the diode and the inductors 250,227 at the end of each operating cycle and this flow of current willtend to reset the saturable inductor 250 to forward saturation. In FIG.23B a resistor 237 and a diode 240 are connected from the converteroutput back to the saturable inductor 250 such that current, Ix, canflow from the output and through the inductor 250 in a direction whichresets the inductor to forward saturation during the latter portion ofthe operating cycle when the voltage Vc is below Vout.

The operation of the multi-mode switch 200 of FIG. 17 was explained withreference to the magnetization characteristic of FIG. 18. This was donefor purposes of illustration, but it is understood that the principle ofoperation of the switch is not dependent on a specific magnetizationcharacteristic but rather on the impedance variation afforded by thereversion of the saturable inductor between its saturated andunsaturated states. In practice, saturable inductors having a variety ofcharacteristics may be used. For example, FIG. 24 shows a magnetizationcharacteristic for a saturable inductor 350 which exhibits a saturatedinductance, Lsat1, and a relatively smooth increase in inductance ascurrent declines toward zero. The use of distinct saturable and linearinductors 250, 227 in the switch 200 of FIG. 17 is also illustrative. Asingle inductor may also be implemented which has a relatively highunsaturated inductance but which is constructed to exhibit apredetermined saturated inductance corresponding to the desired value ofLr1. For example, the multi-mode switch 300 of FIG. 25 consists of aunidirectional switch 130 in series with a saturable inductor 350 havinga magnetization characteristic like that of FIG. 24 for which, asindicated in FIG. 25, the value of Lsat1 is approximately equal to avalue Lr1, where Lr1 is significantly lower than the unsaturatedinductance. The switch 300 may be applied in the same manner as theswitch 200 of FIG. 17.

A benefit of the multi-mode switch topology 200 of FIG. 17 is that thesaturable inductor 250 provides reverse blocking with significantly lesscircuit loss than a series diode. For example, in a non-isolated ZCSbuck converter operating in half-wave mode, at an input voltage of 5volts, a load voltage of 3.3 volts and at an average load current of 13Amperes, a series Schottky blocking diode (used, as shown in FIG. 5B, inconjunction with an N-channel MOSFET unidirectional switch 26) having arelatively low voltage drop of approximately 0.45 volt exhibits a lossof nearly 6 watts. In an equivalent converter using a saturableinductor, however, the inductor loss is less than 1 Watt--a reduction inoverall loss of more than 5 Watts. Another benefit is that theperformance of the ZCS switch controller can be relaxed. While theperformance of the "ideal" converter of FIG. 1 will depend upon thedegree to which the timing of the turn-off of the switch coincides withthe instant in time at which the current crosses zero, the accuracy ofswitch timing in converters which include series elements for blockingreverse current flow (such as prior art converters using switches of thekind shown in FIGS. 5B, 6B, or converters according to the presentinvention using switches of the kind shown in FIG. 16 or 17) is muchless important. In a prior art half-wave converter (e.g., FIG. 5B) theswitch can be turned off at any time between the first zero crossing ofcurrent and the time at which the voltage Vc declines to a value Vin. Ifthis timing condition is met, the diode 38 will block reverse currentflow and the requirements for zero-current switching will be met. In theswitch of FIG. 17 a similar strategy can be used: the unidirectionalswitch can be turned off at any time after the initial half-wave forwardflow of current is complete but before the saturable inductor saturates.This is relatively easy to accomplish, since, with reference to FIG. 3,the time period between times t1 to t3 is bounded to be betweenTmin=pi,sqrt(Lr1,C) and Tmax=1.5,pi,sqrt(Lr1*C). Thus, one simple way tocontrol the switch would be to sense the beginning of the rise in Vc(t)after the switch is turned-on and then turn the switch off after a fixedtime delay thereafter, the delay being equal, for example, to slightlymore than Tmax. Similar comments apply to switch turn-off following aninterval of reverse current flow. In either case, the inherent currentlimiting property of the saturable inductor allows for simplification ofthe switch controller.

Another benefit of the use of a saturable inductor in a multi-modeswitch is that the minimum converter operating frequency is boundedwithin a range as both load and the input source voltage vary. Withreference to FIGS. 22A and 22B, if we make the simplifying assumptionthat the majority of the area under each waveform occurs during thenon-resonant portion (e.g., between times tw and tz in each Figure) thenthe area under the waveform of FIG. 22A is approximately VSa=2*Vin*Ts1and the area under the waveform of FIG. 22B is approximatelyVSb=2,Vin*Ts2=2,VSa (since, as discussed earlier, Ts2 is approximately2*Ts1). As also noted earlier, however, Vin*Ts1 is set by thevolt-second rating of the saturable inductor so the total area under thewaveform will only vary over an approximately 2:1 range as the load andinput voltage vary. Since the converter output voltage is equal to theaverage value of Vc(t) (assuming no losses in the output filterinductor), and since the average value of Vc(t) is equal to Vout=AREAUNDER Vc(t)/OPERATING PERIOD =fop*AREA, then the operating frequencywill also vary only over a 2:1 range.

The simple dual-mode controller 112 illustrated in FIGS. 11A and 11Bdoes not provide the bounded variation in lower frequency limit affordedby the use of a saturable inductor. Because the controller of FIG. 11incorporates a fixed delay time (e.g., delay time Td, FIG. 12D), aconverter using such a controller will exhibit a range of variation ofoperating frequency which is also dependent upon input voltage. As aresult, the total range of variation will be approximately two times theanticipated variation in the value of Vin. However, a dual-modecontroller can be arranged in a virtually infinite number of ways toprovide for some desired behavior in converter minimum operatingfrequency. For example, the dual-mode controller 112a of FIG. 11C isarranged to emulate an ideal saturable inductor when used in amulti-mode switch of the kind shown in FIG. 11A. In FIG. 11C, eachincoming trigger pulse 77 initiates a half-wave forward transfer phasevia ZCS controller 12a. As Vc(t) rings up above Vin, the integrator isenabled and begins to deliver an output, VS(t), proportional to the timeintegral of Vc(t). If Vc(t) remains above Vin for a period of timesufficient for VS(t) to exceed some predetermined value, VS (equivalentto a volt-second rating), then the output of comparator 382 will gohigh, triggering monostable 76 and beginning a reverse energy transferphase via ZCS controller 12b. If, however, the load is sufficiently highso that Vc(t) drops below Vin prior to VS(t) becoming greater than VS,then the output of comparator 70 will go low, the integrator 380 will bedisabled and its output reset to zero, and the reverse energy transferphase will not occur.

While the preceding description has focused on a limited number ofimplementations of a ZCS buck converter, the scheme can be applied toany ZCS converter topology. For example, FIG. 26A shows a prior art ZCSboost converter 370; FIG. 26B shows waveforms for the voltage Vc(t) inthe converter, when operated in the half-wave mode, as the load isreduced from the maximum allowable load value, Pmax, to lower values,Pi>P2>P3, at a constant output voltage Vout>Vin. Like the half-wave ZCSbuck converter, the operating frequency of the boost converter 370 willdecline essentially linearly with load.

FIG. 27A shows a dual-mode ZCS boost converter 470 which corresponds tothe topology of FIG. 26A but which is modified to include a multi-modeswitch 400. The switch can be of the kind shown in FIGS. 11, 14 or16A-16D, incorporating a dual-mode switch controller (not shown, but ofthe kind previously described) for controlling the forward and reverseflow of switch current, Isw, or it can be a switch of the kind shown inFIGS. 17 and 25, using a saturable inductor. At relatively high valuesof load the converter 470 will operate in a half-wave mode: the switch400 will be opened upon the first zero-crossing of current, Isw,following switch closure and only positive Isw will flow. The voltageVc(t) will appear as shown in FIG. 26B. Below some value of load,however, the switch 400 will be closed again at a time subsequent to theend of the initial half-wave flow of current provided that the voltage,Vc, is below zero volts. This is illustrated in FIGS. 27B, 27C and 27D.In the Figures the operating cycle begins with closure of the switch 400at time t=to. Between times t=to and t=t1, the current Isw ramps up(FIG. 27C) as the current Io ramps down (FIG. 27D). The positivehalf-wave flow of current, Isw, ends at time t2, after which a reverseflow of Isw is blocked by the switch 400. At a time t3>t2, the voltageVc(t2) (FIG. 27B) is below zero volts and the switch is closed again,causing a reverse flow of current, Isw, and reverse transfer of energyback to the input source 14, between times t3 and t4. This isaccompanied by a symmetrical "ringup" of the voltage Vc about zerovolts, from a value Vc(t3)=-Vx to a value Vc(t4)=+Vx. At time t=t4positive flow of current Isw is blocked by the switch 400. Followingtime t=t4, the voltage Vc will continue to ramp up (at a ratedVc/dt=Iin/Cr) until it reaches a value Vout at time t=t5, at whichpoint current, Io, will once again flow into the load 18. A dual-modeZCS boost converter will provide the same benefits previously describedfor the dual-mode ZCS buck configuration: by setting a lower bound onconverter operating frequency it allows for a smaller, more efficient,converter; in configurations using a saturable inductor in themulti-mode switch, the full efficiency potential of half-wave ZCSconversion, at elevated loads, may be more closely achieved.

The invention is also applicable to ZCS converters which operatedirectly from bipolar input sources. One example of such a converter, ofthe kind described in Vinciarelli, et al, "AC to DC Boost PowerConverters," U.S. patent application Ser. No. 08/274,991, is shown inFIG. 28. In the Figure the voltage, Vin, delivered by the bipolar inputsource 214 may assume either positive or negative values. In thehalf-wave operating mode, a bipolar switch 430 (e.g., one which iscapable of blocking a bipolar voltage, Vsw, when open, and capable ofconducting a bipolar current, Isw, when closed) is opened at the firstzero-crossing of the current Isw following closure of the switch,resulting in a capacitor voltage waveform similar to that shown in FIG.26B (with the exception that the polarity of the waveform will reversefor negative values of Vin).

FIG. 29 shows a dual-mode ZCS AC to DC boost converter 570 whichcorresponds to the topology of FIG. 28 but which is modified to includea multi-mode switch 500. Since the multi-mode switch 500 must be abipolar switch, multi-mode switches of the kind shown in FIGS. 11, 14and 16A-16D may be used, but switches of the kind shown in FIGS. 17 and25 may not. In application, switches of the kind shown in FIGS. 11, 16Aand 16C provide no apparent benefit over those in FIGS. 14, 16B and 16D,since the values of the two inductors 27a, 27b must be set to the samevalue (because the roles of the two switches 130a, 130b, will reversewith the polarity of the source in terms of whether they carry energyforward from the input source toward the load or in reverse back fromthe capacitor 442 to the input source). Where it is desirable to providedifferent characteristic time constants for the forward and reverseenergy transfer phases, two bipolar multi-mode switches (e.g., of thekind shown in FIG. 16B) comprising inductors of different values, may beparalleled and controlled in accordance with the polarity of the inputsource to. Bipolar multi-mode switches of the kind shown in FIGS. 30Aand 30B, incorporating unidirectional switches 130a, 130b and diodes434a, 434b, 438a, 438b in combination with either a saturable inductor450 alone, or a saturable inductor in series with a linear inductor 427,may also be used in bipolar ZCS dual-mode converters. While thesaturable inductors cannot replace the series blocking diodes in abipolar ZCS dual-mode converter, their volt-second rating can be used toprovide the "timing" of the automatic initiation of the reverse energytransfer phase as load is reduced.

Use of dual-mode control may also be used in galvanically isolated ZCSconverters. For example, FIG. 31 shows an isolated half-wave ZCS forwardconverter 480 of the kind described in Vinciarelli, "Forward ConverterSwitching at Zero-Current," U.S. Pat. No. 4,415,959, November, 1983(incorporated by reference). The converter includes a circuit 490 forresetting the magnetic core of the transformer of the kind described inVinciarelli, "Optimal Resetting of the Transformer's Core in SingleEnded Forward Converters," U.S. Pat. No. 4,441,146, April, 1984(incorporated by reference). In operation the characteristic timeconstant for the converter is set by the value of the transformer 464leakage inductance, L1, and the value of the capacitor 466, Cr. The mainswitch 460 is turned on and off at times when the forward current, Iin,is zero, and the reset switch 462 is controlled to be on when the mainswitch is off and to turn off prior to the main switch turning on. Adual-mode version of the ZCS converter 492 of FIG. 31A is shown in FIG.31B. In the Figure the diode of FIG. 31A has been replaced with a secondswitch 469. A dual-mode controller 112 (e.g., of the kind shown in FIGS.11A and 11B) turns the main switch 460 and the second switch 469 on andoff simultaneously. The reset switch 490 is controlled as in the priorart converter: it is turned on when the main switch is off and is turnedoff just prior to the main switch turning on. In the converter of FIG.31B, all of the switches 460, 469, 490 can be embodied as unidirectionalswitches, as is illustrated using MOSFETs in the partial schematic inFIG. 32.

In general, a dual-mode ZCS power converter would typically include: (1)a switching element combined with circuit elements (e.g., inductors,capacitors), the circuit elements serving to both define thecharacteristic time constant(s) for the rise and fall of the currentsand voltages in the converter during each converter operating cycle andarranged so that the switching element may be opened and closed at timesof zero current, and (2) means for performing dual-mode control of theswitching element. Dual-mode control provides for two distinct operatingmodes. In a first operating mode a switching element is closed andopened once during each operating cycle, at consecutive times of zerocurrent, to initiate a forward transfer of energy from the input sourcetoward the load; in a second operating mode, a first closing and openingof a switching element, at consecutive times of zero current during afirst phase of the operating cycle, associated with initiation offorward energy transfer from the input source toward the load, isfollowed a finite time later by a second phase during which a switchingelement is once again closed and opened at consecutive times of zerocurrent, provided that operating conditions within the converter willallow for a reverse transfer of energy back to the input source.

The multi-mode switch may, as illustrated in FIGS. 33A and 33B, begeneralized into a family of multi-mode resonant switches. In accordancewith the terminology used by Liu, et al, in "Resonant Switches--AUnified Approach to Improve Performances of Switching Converters," IEEEInternational Telecommunications Conference, 1984 Proceedings, pp.344-351 (incorporated by reference), to describe different types ofprior art resonant switches, the dual-mode resonant switch of FIG. 33Ais referred to as an M-Type dual-mode resonant switch 403 and thedual-mode resonant switch of FIG. 33B is referred to as an L-Typedual-mode resonant switch 409. Both the M-Type and the L-Type dual-moderesonant switches include a multi-mode switch 407 and a capacitor 405.The multi-mode switch 407 can be any of the ones shown in FIGS. 11, 14or 16A-16D, incorporating a dual-mode switch controller (not shown, butof the kind previously described) for controlling the forward andreverse flow of switch current, Isw, or it can be a switch of the kindshown in FIGS. 17, 25, 30A and 30B, using a saturable inductor. Ingeneral, inductance included in series with the switching element withinthe multi-mode switch 407 provides for closing of the switching elementat times of zero current and a limit on the rate-of-change of switchcurrent thereafter; the combination of the inductance and thecapacitance 405 causes a sinusoidal time variation in the switchcurrent, thereby providing a zero-crossing of current, subsequent toturn-on, at which to turn the switching element off. The dual-modeswitch controller provides for two distinct operating modes for thedual-mode resonant switches 403, 409. In a first operating mode aswitching element within the multi-mode switch 407 is closed and openedonce during each operating cycle, at consecutive times of zero current,to initiate a flow of current, Isw, of a particular polarity; in asecond operating mode, the first closing and opening of a switchingelement, at consecutive times of zero current during a first phase ofthe operating cycle associated with initiation of a flow of Isw at aparticular polarity, is followed a finite time later by a second phaseduring which a switching element in the multi-mode switch 407 is closedand opened at consecutive times of zero current, provided that thesecond phase will allow for a flow of current Isw of an oppositepolarity.

Other embodiments are within the scope of the following claims. As ageneral rule, for example, any ZCS power conversion topology which canbe operated in the full-wave operating mode can be adapted to dual-modecontrol. Thus, any of the wide variety of full-wave ZCS convertertopologies which are described in Lee, et al, "Zero-Current SwitchingQuasi-Resonant Converters Operating in a Full-Wave Mode," U.S. Pat. No.4,720,667, Jan. 19, 1988 (incorporated by reference), including the buck(Lee, FIGS. 11A through 11F), boost (Lee, FIGS. 12A through 12F),buck-boost (Lee, FIGS. 14A through 14P), forward (Lee, FIGS. 15B and15C), flyback (Lee, FIGS. 16B and 16C), Cuk (Lee, FIGS. 17C, 17D, 18Band 18C) and SEPIC topologies (Lee, FIGS. 19B and 19C), and those whichare also described in Vinciarelli, et al, "Boost Switching PowerConversion," U.S. Pat. No. 5,321,348, Jun. 14, 1994 (incorporated byreference) may be adapted to dual-mode control, and each of thesetopologies (e.g., as shown in the referenced figures in Lee) willtherefore have a corresponding dual-mode ZCS topology in accordance withthe present invention (e.g., in each of the referenced figures in Leethe switch labeled S1 would be replaced with an appropriate multi-modeswitch). The time delay between the initiation of half-wave forwardcurrent flow at the beginning of an operating cycle in a dual-mode ZCSconverter, or in a dual-mode resonant switch, and the initiation of areverse flow of current later in the cycle need not be a fixed delay butcan be made variable to accommodate some particular application need orconverter operating requirement.

What is claimed is:
 1. A method for controlling the minimum operatingfrequency of a zero-current switching converter which transfers powerfrom an input source for delivery to a time-varying load by opening andclosing a switch at times of zero current flow, comprisingin a firstportion of each of a series of converter operating cycles, controllingsaid switch to initiate forward energy transfer from said input source,and in response to changes of said load, during each of some of theconverter operating cycles and at times which are not contiguous withsaid first portion, controlling said switch to initiate reverse energytransfer to said input source.
 2. The method of claim 1 furthercomprisingcausing a fixed time delay between the beginning of said firstportion and the initiation of said reverse energy transfer.
 3. Themethod of claim 1 further comprizingselectively adjusting the time delaybetween the beginning of said first portion and the initiation of saidreverse energy transfer.
 4. The method of claim 3 furthercomprisinginitiating the reverse energy transfer on the basis of avoltage measured across a capacitor.
 5. The method of claim 1 furthercomprisingadapting the volt-second characteristic of a saturableinductor to effect a &ontrolled time delay between the beginning of saidfirst portion and the initiation of said reverse energy transfer.
 6. Themethod of claim 1 further comprisinginitiating the reverse energytransfer if doing so will result in energy being transferred from saidconverter back to said input source.